High range resolution radar system

ABSTRACT

Described herein is a method and apparatus for improving high range resolution of a radar system. The method comprises phase shifting a radar pulse to be transmitted at substantially the radar transmission frequency and phase shifting the received radar pulse at substantially the radar transmission frequency. The phase shifting is implemented using a monolithic microwave integrated circuit (MMIC) ( 42 ) driven by a digital circuit ( 44 ) to provide a phase profile which is applied to radar pulse ( 52 ) produced by a radar pulse generator ( 54 ) and which is also applied to a received radar pulse ( 60 ). A master clock and synchronizer ( 72 ) provides clock signals for an analogue to digital converter (ADC) ( 68 ), the generator ( 54 ) and the digital circuit ( 44 ) so that the MMIC ( 42 ) is clocked at a frequency which is directly harmonically related to the ADC ( 68 ). This avoids spurious beat frequencies which could interfere with a wanted radar signal.

The present invention relates to improvements in or relating to rangeresolution, and is more particularly concerned with obtaining high rangeresolution for a radar system for example.

In order to obtain high range resolution, a radar must transmit andprocess a wide bandwidth waveform. For example, to obtain a rangeresolution of 1 m, a nominal bandwidth of 150 MHz (in practice somewhatmore) must be used.

Conventional radar receivers operate with much narrower bandwidths,typically in the range 1˜10 MHz. Whilst specialised wideband receiverscan be used, they are relatively difficult to implement and are costly,especially in multifunction radar.

A more efficient method of obtaining high range resolution is known as‘stretch radar’ or deramp processing. The conventional approach is touse a voltage controlled oscillator to provide the frequency modulatedsignals on transmit and receive. On receive, the signal feeds thereceiver's local oscillator.

A wideband pulse is transmitted where the carrier frequency is rampedlinearly over the pulse duration, the total frequency excursion beingthe transmitted bandwidth. At the expected time of reception from adistant target, the radar receiver's local oscillator is ramped at thesame rate. This has the result that the downconverted signal is at zerofrequency if the deramp is precisely synchronised with the receivedpulse, or has a small (constant) frequency offset if the pulse is notprecisely aligned. This frequency offset may be extracted by Fourieranalysis of the received pulse. With this method a much finer rangeresolution is achieved than would otherwise be possible with anarrowband receiver, provided the parameters are chosen appropriately.However, this method only works over a limited range swathe which iseffectively delimited at one end by the start of the deramp, and at theother by the point at which the frequency offset signal goes outside thebandwidth of the narrowband receiver. This limitation is rarely aproblem in practice, as high resolution radars are normally onlyinterested in small regions.

Whilst the method described above is based on frequency modulation andemploys variable frequency oscillators, its implementation is relativelycomplex and suffers from the disadvantage that the circuits employedsuffer from drift and other errors or requires complex calibrations ifdirect digital synthesis is utilised.

It is therefore an object of the present invention to provide a methodof obtaining high range resolution which overcomes the problemsdescribed above.

In accordance with one aspect of the present invention, there isprovided a method of obtaining high range resolution in a radar system,the method comprising the steps of:

a) generating a radar pulse;

b) modulating the radar pulse;

c) transmitting the modulated radar pulse;

d) receiving a radar pulse;

e) modulating the received radar pulse; and

f) processing the modulated radar pulse to obtain range information;

characterised is that step b) comprises applying a time-dependent phaseshift, changed at discrete time intervals, to the radar pulse atsubstantially the radar transmission frequency, and step e) comprisesapplying a time-dependent phase shift, changed at discrete timeintervals, to the received radar pulse at substantially the radartransmission frequency;

and in that step f) comprises sampling the received signal at discretetime intervals which are an integral number of the time intervals of thetime-dependent phase shift.

By the term “at substantially the radar transmission frequency” isintended to mean that the phase shift applied is substantially at thetransmission frequency. For example, if the radar transmission frequencyis 10 GHz, the radar pulse is at 1 GHz, then the phase shift is appliedat a frequency of 9 GHz which is substantially the same as thetransmission frequency.

The time-dependent phase shift may comprise a synthesised sequence or anarbitrary sequence.

Ideally, the time-dependent phase shift produces a predetermined phaseprofile and it is preferred that such a phase profile comprises aquadratic phase profile.

In accordance with the second aspect of present invention, there isprovided a radar system comprising:

means for generating a radar pulse;

means for modulating the radar pulse;

means for transmitting the radar pulse;

means for receiving a reflected radar pulse;

means for modulating the received radar pulse; and

means for processing the modulated received radar pulse to obtain rangeinformation;

characterised in that the means for modulating the radar pulse includesa phase shifter which applies a time-dependent phase shift, which ischanged at discrete time intervals, at substantially the radartransmission frequency, and the means for modulating the received radarpulse includes a phase shifter which applies a time-dependent phaseshift, which is changed at discrete intervals, at substantially theradar transmission frequency;

and in that the means for processing the modulated received radar pulseincludes sampling means for sampling the received signal at discretetime intervals which are an integral number of the time intervals of thetime-dependent phase shift.

Each phase shifter may be driven in accordance with a synthesisedsequence which may be implemented by a plurality of discrete logiccomponents or by a field programmable gate array.

Alternatively, each phase shifter may be driven in accordance with anarbitrary sequence provided by a memory device.

If is preferred that the means for modulating the radar pulse includes alocal oscillator and first mixing means, the local oscillator providinga signal for mixing with the radar pulse in the first mixing means. Thelocal oscillator signal may be phase shifted prior to mixing with theradar pulse. Alternatively, the local oscillator signal may be mixedwith the radar pulse prior to being phase shifted.

Similarly, it is preferred that the means for modulating the receivedradar pulse includes a local oscillator and a second mixing means, thelocal oscillator providing a signal for mixing with the received radarpulse in the second mixing means. The local oscillator signal may bephase shifted prior to mixing with the received radar pulse.Alternatively, the received radar pulse may be phase shifted prior tomixing with the local oscillator signal.

In one embodiment of the present invention, a single local oscillator isutilised which provides a local oscillator signal to both the radarpulse and the received radar pulse. In this embodiment a single phaseshifter is located in the local oscillator signal path and the samephase shifted signal is utilised for modulation of both the radar pulseand the received radar pulse, the modulation being applied at differenttimes.

Ideally, each phase shifter comprises a digital phase shifter, forexample, a monolithic microwave integrated circuit.

For a better understanding of the present invention, reference will nowbe made, by way of example only, to the accompanying drawings in which:

FIG. 1 illustrates a frequency-time diagram for stretch radar;

FIG. 2 illustrates a phase shifter operation;

FIG. 3 illustrates a block diagram of a device for implementing stretchradar in accordance with the present invention;

FIG. 4 illustrates a block diagram illustrating a first implementationof modulation of the radar pulse and the received radar pulse; and

FIGS. 5 and 6 each illustrates a further implementation of modulation ofthe radar pulse and the received radar pulse.

As background, stretch radar decoding is described. For a narrow rangeswathe, such as is mapped by a synthetic aperture radar, linearfrequency modulation is commonly decoded by a technique called stretchradar. As the return from the swathe is received, its frequency issubtracted from a reference frequency that increases at the same rate asthe transmitter frequency. The reference frequency increasescontinuously throughout the entire period in which the return from theswathe is received. Consequently, the difference between the referencefrequency and the frequency of the return from any one point on theground is constant. If the initial offset, f₀, of the referencefrequency is subtracted from the difference obtained, the result isproportional to the range of the point from the near edge of the swathe.Range is thus converted to frequency.

By considering four closely spaced points after subtraction has beenperformed, four returns are received which, although received almostsimultaneously, have slightly staggered arrival times. Due to thedifference or stagger in arrival times discernible differences infrequency are obtained for each of the four returns. These differencesin frequency can then be converted to range values. This is achieved byapplying the output of a synchronous detector to a bank of narrowbandfilters implemented, for example, with a highly efficient fast Fouriertransform.

Turning now to FIG. 1, a transmitted pulse 10 is shown which has a totalfrequency excursion B and a transmitted pulse width T_(p). The pulse ata ‘target’ 12 is also shown. Two received pulses 14, 16 are also shownrelative to a reference pulse 18. The carrier frequency, f₀, isrepresented by chain line 20. As shown, at a point in time, thefrequency difference between the reference pulse 18 and the firstreceived pulse 14 is Δf and for a given frequency, the time delay is Δt.

If the minimum target range to start of range swathe is R_(min), it canbe expressed as

$R_{\min} = {T_{p}\frac{c}{2}}$where c is the speed of light.

For a transmitted pulse width of 100 μs, R_(min) is 15 km.

If the range swathe is R_(s), the time difference, T_(s), between pointson nearest and furthest boundaries of the swathe can be expressed as

$T_{s} = {2\frac{R_{s}}{c}}$Therefore, the total time duration for the ‘de-stretch’ must beT_(p)+T_(s). For example, if R_(s) is 1 km, T_(s) is 6.667 μs.

Range resolution, R_(r), can be expressed as:

$R_{r} = {\left( {F_{r}\frac{c}{2}} \right)\left( \frac{T_{p}}{B} \right)}$where F_(r) is the minimum discernible frequency resolution.

Then,

$B = {\left( {F_{r}\frac{c}{2}} \right)\left( \frac{T_{p}}{R_{r}} \right)}$If, for example, F_(r) is 10 kHz, R_(r) is 1 m, T_(p) is 100 μs, then Bis 150 MHz, and the slope k of the frequency sweep is 1.5 MHz per μs

$\left( {k = \frac{B}{T_{p}}} \right).$

A digital phase shifter is operated to sweep the frequency by ±B/2 aboutthe carrier frequency, f₀, that is, the total frequency excursion B.However, to satisfy the Nyquist criterion, the clock rate of the digitalphase shifter would have to be in excess of B. For example, if B is 200MHz, the clock rate will need to be greater than 200 MHz.

If the ‘de-stretch’ is centred on the centre of the range swathe, thenthe intermediate frequency (IF) bandwidth is symmetrical about the IFcentre frequency. The IF bandwidth, B_(IF), can be expressed as beingapproximately

$B_{IF} = {k\left( \frac{2R_{s}}{c} \right)}$As an example, if k=1.5 MHz per μs, as in the example above, and R_(s)is 1 km, the B_(IF) is 1 MHz. If the ‘de-stretch’ commences at the startof the range swathe, then the IF bandwidth is to one side of the IFcentre frequency.

Referring now to FIG. 2 for a description of phase shifter operation, afrequency generator 30 is shown connected to a first phase shifter 32.The frequency generator 30 provides an output signal cos(ω₀t). The firstphase shifter 32 is connected to a second phase shifter 34 via a delayline 36. The delay line 36 produces a delay, T, which represents thedelay to point target within range swathe.

If the output, A(t), of the first phase shifter 32 is to have afrequency ω, whereω=ω₀+2πktthat is, the linear frequency sweep of slope k, then

${A(t)} = {\cos\left( {{\omega_{0}t} + \frac{2\pi\;{kt}^{2}}{2}} \right)}$where the phase shift

${\theta_{A}(t)}\mspace{14mu}{is}\mspace{14mu}{\frac{2\pi\;{kt}^{2}}{2}.}$

After the transmission delay, T, in delay line 36,

${A\left( {t - T} \right)} = {\cos\left( {{\omega_{0}\left\lbrack {t - T} \right\rbrack} + \left( \frac{2\pi\;{k\left\lbrack {t - T} \right\rbrack}^{2}}{2} \right)} \right)}$

The second phase shifter 34 has to be set to produce a phase shift,θ_(B)(t), of

${\theta_{B}(t)} = \frac{{- 2}\pi\;{k\left( {t - T_{0}} \right)}^{2}}{2}$which has the opposite characteristic to the first phase shifter 32,that is, the slope is negative instead of being positive. Time T₀represents the time to the start or centre of the range swathe asappropriate.

The output, B(t), of the second phase shifter 34 is then

$\begin{matrix}{{B(t)} = {\cos\left( {{\omega_{0}\left\lbrack {t - T} \right\rbrack} + \left( \frac{2\pi\;{k\left\lbrack {t - T} \right\rbrack}^{2}}{2} \right) - \left( \frac{2\pi\;{k\left( {t - T_{0}} \right)}^{2}}{2} \right)} \right)}} \\{= {\cos\left( {{\omega_{0}t} + {\frac{2\pi\; k}{2}\left( {T^{2} - {2{Tt}} - T_{0}^{2} + {2T_{0}t}} \right)} - {\omega_{0}T}} \right)}} \\{= {\cos\left( {{\left\lbrack {\omega_{0} - {2\pi\;{k\left( {T - T_{0}} \right)}}} \right\rbrack t} - {\omega_{0}T} - {\frac{2\pi\; k}{2}\left\lbrack {T_{0}^{2} - T^{2}} \right\rbrack}} \right)}}\end{matrix}$

It will be appreciated that the above expression includes both afrequency or time dependent term and a static phase term, and thefrequency offset, Δf, from the reference pulse 18 (FIG. 1) isΔf=−k(T−T ₀)

In practice, both phase shifters 32, 34 will be digitally controlleddevices having a finite number of phase states, usually 2^(m), and thecontrol data will be clocked out once every T_(c) seconds where

$T_{c} = \frac{1}{f_{c}}$where f_(c) is the frequency at which the phase shifters are clocked.

The angular frequency, ω_(p), from a phase shifter is given by

$\begin{matrix}\; & {\omega_{p} = \frac{\mathbb{d}\theta}{\mathbb{d}t}} \\{but} & \; \\\; & {{d\;\theta} = {p\left( \frac{2\pi}{2^{m}} \right)}} \\{where} & \; \\\; & {{p = 0},1,2,{\ldots\mspace{11mu} 2^{m}}} \\{and} & \; \\\; & {{dt} = {qT}_{c}} \\{where} & \; \\\; & {{q = 1},2,3,\ldots} \\{{Therefore},} & \; \\\; & {\omega_{p} = {\frac{\left\lbrack {p\left( \frac{2\pi}{2^{m}} \right)} \right\rbrack}{{qT}_{c}} = \frac{2\pi\;{pf}_{c}}{q\left( 2^{m} \right)}}}\end{matrix}$

To satisfy the Nyquist criteria, the highest frequency, ω_(pmax), whichcan be produced is

ω_(p  max ) = π f_(c) $f_{p\mspace{11mu}\max} = \frac{f_{c}}{2}$which means that p=2^(m−1) and q=1.

Before describing FIG. 3, it is to be noted that frequency can bethought of in terms of phase shifting as frequency is simply a rate ofchange of phase. This is relevant to the operation of the device in FIG.3. As a result of frequency being considered in terms of rate of changeof phase, the linear frequency ramps required for stretch radar may besimple quadratic phase profiles.

Moreover, phase shifting devices operating directly at radio frequency(RF) may be realised in gallium arsenide monolithic microwave integratedcircuits (GaAs MMICs).

Turning now to FIG. 3, a device 40 for implementing stretch radar isshown. The device 40 comprises a MMIC phase shifter 42 which is drivenby a field programmable gate array (FPGA) drive circuit 44. The FPGAdrive circuit 44 produces a synthesised sequence which drives the phaseshifter 42. As an alternative to a FPGA drive circuit 44, thesynthesised sequence can be implemented by discrete logic components. Inanother embodiment, the phase shifter 42 may be driven by an arbitrarysequence stored in a suitable memory device (not shown). The phaseshifter 42 is connected to a radar RF reference oscillator 46 andprovides a local oscillator (LO) output signal 48 which is passed to amixer 50 where it is mixed with a pulse 52 from a radar pulse generator54. The mixer 50 produces a phase shifted (upconverted) pulse 56 whichis passed to a radar transmitter (not shown).

It is important that the phase shifter is applied to a signal which hasa frequency which is substantially at the transmission frequency. Forexample, the radar pulse generator may generate a pulse at a frequencyof around 1 GHz and the local oscillator signal is at a frequency ofaround 9 GHz, and when the pulse and the oscillator signals are mixed, atransmission frequency of around 10 GHz.

The LO output signal 48 is also passed to a mixer 58 where it is mixedwith an incoming radar pulse 60 from a radar antenna (not shown). Themixer 58 produces a (downconverted) pulse 62 which has the LO signal 48subtracted from it. The pulse 62 is then passed to a receiver 64. Thereceiver 64 provides an output signal 66 which fed to ananalogue-to-digital converter (ADC) 68, which provides a digital signal70 for passing to a radar processor (not shown). A radar master clockand synchroniser 72 is connected to the ADC 68 to provide an ADC clocksignal 74. The synchroniser 72 also provides a clock signal 76 for thedrive circuit 44 which harmonically clocks the drive circuit 44 inrelation to the ADC clock signal 74.

In operation, the upconverted pulse 56 is transmitted to a scene (notshown) by a transmitter (also not shown). A reflected or returned pulse(not shown) from an object (not shown) in the scene is received at theantenna (also not shown) and the pulse 60 from the antenna isdownconverted in mixer 58 to provide pulse 62, pulse 62 being processedto provide the required range information of the object in the scene.

It will readily be understood that a fixed frequency transmit pulse 52is combined with a phase shifted signal 48 derived from a MMIC phaseshifter 42 which is driven by circuit 44 to predetermined phase profile,for example, a quadratic phase profile. Substantially the same signal 48is used to deramp the received pulse. The resulting signal is thenprocessed in the normal way. In the embodiment described with referenceto FIG. 3, the phase shifter 42 is in the local oscillator path andhence the same phase shift is applied to transmitted and receivedsignals.

As described above with reference to FIG. 3, a single MMIC phase shifter42 is used for both upconversion and downconversion. It will beunderstood that separate phase shifters, and associated drivingcircuits, can be used for the upconversion and the downconversion. It isan essential feature of the invention that the MMIC phase shifters areclocked at a frequency directly harmonically related to the radar'sanalogue-to-digital converters. This is vital in order to avoid spuriousbeat frequencies which are difficult to control and which could damagethe wanted radar signal.

The device of the present invention has the following advantages:

-   1. The implementation is totally digital and requires no setup or    calibration.-   2. The implementation is stable and requires no adjustment.-   3. The circuitry required is extremely compact and low cost.-   4. The technique may be easily retrofitted to a wide range of    existing radars, providing a substantial improvement in performance    at low cost.

FIGS. 4 to 6 illustrate different implementations of the localoscillator signal and phase shifting for both transmission andreceiving. Again, as before, in each implementation, the circuitry maybe designed to accommodate either two phase shifters, one in each of thetransmission and receiving paths, or a single phase shifter whichoperates in both the transmission and receiving paths, and will dependon the particular application.

In FIG. 4, a first alternative implementation is shown which isgenerally designated as 80. A radar pulse 82 is mixed with a localoscillator (LO) signal from LO 84 in mixer 86. The output from the mixer86 is then passed to a phase shifter 88 prior to transmission astransmission signal 90. On receiving, signal 92 from an antenna (notshown) is mixed with a LO signal which has been phase shifted by phaseshifter 94 in mixer 96 to provide a received signal 98 for furtherprocessing. In this implementation, both phase shifters 88 and 94 applythe same phase shift (e.g. both positive) to both the outgoing andincoming signals.

In a second alternative implementation as shown in FIG. 5 and designated100, components which have been previously described in FIG. 4 arereferenced alike. In this implementation, the radar pulse 82 is mixedwith a phase shifted LO signal in mixer 86 to produce signal 90 forsubsequent transmission. The signal 92 received from the antenna (notshown) is applied to the phase shifter 94 before being downconverted bymixing with the LO signal in mixer 96 to produce the received signal 98for further processing (not shown). Here, as the phase shifter 94 is inthe antenna path, a phase shift opposite to that applied to the LOsignal in the transmission path. For example, phase shifter 88 applies apositive phase shift and phase shifter 94 applies a negative phaseshift.

In a third alternative implementation as shown in FIG. 6 and designated110, components which have been previously described in FIG. 4 arereferenced alike. In this implementation, the radar pulse 82 is mixedwith the LO signal in mixer 86 prior to passing to the phase shifter 88to produce the transmission signal 90. On the incoming path, signal 92received from the antenna (not shown) is applied to phase shifter 94prior to being mixed with the LO signal in mixer 96 to produce thereceived signal 98 which is subsequently processed. As with the FIG. 5implementation, the phase shifter 94 is in the antenna path and thephase shift applied is opposite to that of the phase shift applied byphase shifter 88 in the transmission path.

When it is stated that the applied phase shift is opposite, it is meantthat the sign of the phase shift is opposite but the magnitude etc. ofthe phase shift is the same.

Although the present invention has been described above using digitalphase shifters it will be appreciated that analogue phase shifters couldalso be used in accordance with a particular application. However,analogue phase shifters tend to be less stable than digital equivalentsand produce less repeatable output signals than can be achieved withdigital phase shifters.

In all the embodiments or implementations described above, it isimportant that the phase shifters be harmonically related to theanalogue-to-digital clock signal applied to the received signal forsubsequent processing as described with reference to FIG. 3 above.

1. A method of obtaining high range resolution in a radar system, themethod comprising the steps of: a) generating a radar pulse; b)modulating the radar pulse; c) transmitting the modulated radar pulse;d) receiving a radar pulse; e) modulating the received radar pulse; andf) processing the modulated radar pulse to obtain range information;characterized in that step b) comprises applying a time-dependent phaseshift, changed at discrete time intervals, to the radar pulse atsubstantially the radar transmission frequency, and step e) comprisesapplying a time-dependent phase shift, changed at discrete timeintervals, to the received radar pulse at substantially the radartransmission frequency; and in that step f) comprises sampling thereceived signal at discrete time intervals which are an integral numberof the time intervals of the time-dependent phase shift.
 2. A methodaccording to claim 1, wherein the time-dependent phase shift comprises asynthesized sequence.
 3. A method according to claim 1, wherein thetime-dependent phase shift comprises an arbitrary sequence.
 4. A methodaccording to claim 1, wherein the time-dependent phase shift produces apredetermined phase profile.
 5. A method according to claim 4, whereinthe predetermined phase profile comprises a quadratic phase profile. 6.A radar system comprising: means for generating a radar pulse; means formodulating the radar pulse; means for transmitting the radar pulse;means for receiving a reflected radar pulse; means for modulating thereceived radar pulse; and means for processing the modulated receivedradar pulse to obtain range information; characterized in that the meansfor modulating the radar pulse includes a phase shifter which applies atime-dependent phase shift, which is changed at discrete time intervals,at substantially the radar transmission frequency, and the means formodulating the received radar pulse includes a phase shifter whichapplies a time-dependent phase shift, which is changed at discreteintervals, at substantially the radar transmission frequency; and inthat the means for processing the modulated received radar pulseincludes sampling means for sampling the received signal at discretetime intervals which are an integral number of the time intervals of thetime-dependent phase shift.
 7. A system according to claim 6, whereineach phase shifter is driven in accordance with a synthesized sequence.8. A system according to claim 7, wherein the synthesized sequence isimplemented by a plurality of discrete logic components.
 9. A systemaccording to claim 7, wherein the synthesized sequence is implemented bya field programmable gate array.
 10. A system according to claim 6,wherein each phase shifter is driven in accordance with an arbitrarysequence which is provided by a memory device.
 11. A system according toclaim 6, wherein the means for modulating the radar pulse includes alocal oscillator and first mixing means, the local oscillator providinga signal for mixing with the radar pulse in the first mixing means. 12.A system according to claim 11, wherein the local oscillator signal isphase shifted prior to mixing with the radar pulse.
 13. A systemaccording to claim 11, wherein the local oscillator signal is mixed withthe radar pulse prior to being phase shifted.
 14. A system according toclaim 11, wherein the means for modulating the received radar pulseincludes a local oscillator and second mixing means, the localoscillator providing a signal for mixing with the received radar pulsein the second mixing means.
 15. A system according to claim 14, whereinthe local oscillator signal is phase shifted prior to mixing with thereceived radar pulse.
 16. A system according to claim 14, wherein thereceived radar pulse is phase shifted prior to mixing with the localoscillator signal.
 17. A system according to claim 11, wherein a singlelocal oscillator is utilized which provides a local oscillator signal toboth the radar pulse and the received radar pulse.
 18. A systemaccording to claim 17, wherein a single phase shifter is utilized forboth modulation of the radar pulse and modulation of the received radarpulse.
 19. A system according to claim 6 wherein each phase shiftercomprises a digital phase shifter.
 20. A system according to claim 19,wherein the digital phase shifter comprises a monolithic microwaveintegrated circuit.